Drive device, method thereof, and wireless power transmission device

ABSTRACT

According to one embodiment, a drive device drives “N” number (N is an integer of “2” or greater) of inverters to generate AC power and transmit respective AC power to transmission coil units corresponding thereto and includes a switching signal generation circuit. The switching signal generation circuit generates switching signals to drive first to fourth switching elements of each inverter to complementarily drive the first switching element and the second switching element, and complementarily drive the third switching element and the fourth switching element so that a phase difference between an output current of an “M”th (“M” is an integer of 2 or greater and “N” or below) inverter and an output current of an “M−1”th inverter becomes or approach “360×L/N” degrees (“L” is an integer of “1” or greater and less than “N”) and supplies the switching signals to the first to fourth switching elements of the inverters.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of International Application No.PCT/JP2014/065785, filed on Jun. 13, 2014, the entire contents of whichis hereby incorporated by reference.

FIELD

Embodiments described herein relate to a drive device, a method thereof,and a wireless power transmission device.

BACKGROUND

For wireless power transmission, there has been known a method ofconnecting two coils so as to make their generating electromagneticfields opposite. According to the method, the electromagnetic fieldgenerated around the coils can be reduced.

However, when using a plurality of coils, many factors such asinductance of each coil, characteristics of a part connected to eachcoil, characteristics of a counter device to which each coil transmitspower, positional relation between the counter device all need to besymmetrical in the coils, otherwise, the amplitude of the currentflowing in each coil does not become the same. Further, phases of thecurrent flowing through the coils become different and the generatedelectromagnetic field does not become an opposite phase. As a result,effect of reduction in a leaked electromagnetic field is limited.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows an example of a wireless power transmission deviceaccording to the first embodiment;

FIG. 2 shows a relationship between a phase difference and attenuationamount;

FIG. 3 shows another example of the wireless power transmission deviceaccording to the first embodiment;

FIG. 4 shows examples of switching signals and an output waveformaccording to the first embodiment;

FIG. 5 shows an example of a waveform when there is dead time betweenthe switching signals;

FIG. 6 shows other examples of the switching signals and output waveformaccording to the first embodiment;

FIG. 7 shows other examples of the switching signals and output waveformaccording to the first embodiment;

FIG. 8 are examples showing each of when the phase difference of thevoltage waveform is adjusted to 180 degrees and other degrees;

FIG. 9 shows the first example of a switching signal generation circuit;

FIG. 10 shows the second example of the switching signal generationcircuit;

FIG. 11 shows the third example of the switching signal generationcircuit;

FIG. 12 shows an example of another transmission coil unitconfiguration;

FIG. 13 shows a wireless power transmission device including a pluralityof DC power supplies;

FIG. 14 shows another example of the wireless power transmission deviceaccording to the first embodiment;

FIG. 15 shows an example of the wireless power transmission deviceaccording to the second embodiment;

FIG. 16 describes the effect of reduction in an electromagnetic fieldfor a case of three-phases;

FIG. 17 shows an example of the switching signals and output waveformfor a case of three-phases;

FIG. 18 shows an example of the switching signal generation circuit forthe multiphase;

FIG. 19 shows an example of the wireless power transmission deviceaccording to the third embodiment;

FIG. 20 shows another example of the wireless power transmission deviceaccording to the third embodiment;

FIG. 21 shows an example of the wireless power transmission deviceaccording to the fourth embodiment; and

FIG. 22 shows another example of the wireless power transmission deviceaccording to the fourth embodiment.

DETAILED DESCRIPTION

According to one embodiment, a drive device driving “N” number (N is aninteger of “2” or greater) of inverters corresponding to transmissioncoil units includes a switching signal generation circuit.

The inverters each includes a first switching element and a secondswitching element connected together at respective one ends and a thirdswitching element and a fourth switching element connected together atrespective one ends, a connection node of the first and the secondswitching element being connected to one end of each correspondingtransmission coil unit, a connection node of the third and the fourthswitching element being connected to another end of each correspondingtransmission coil unit.

The inverters each generates AC power by driving the first to fourthswitching elements based on a first power-supply voltage supplied toother ends of the first and third switching elements and a secondpower-supply voltage supplied to other ends of the second and fourthswitching elements, and the inverter outputs the AC power to eachcorresponding transmission coil unit.

The switching signal generation circuit generates switching signals todrive the first to fourth switching elements of each inverter tocomplementarily drive the first switching element and the secondswitching element, and complementarily drive the third switching elementand the fourth switching element so that a phase difference between anoutput current of an “M”th (“M” is an integer of 2 or greater and “N” orbelow) inverter and an output current of an “M−1”th inverter becomes orapproach “360×L/N” degrees (“L” is an integer of “1” or greater and lessthan “N”).

The switching signal generation circuit supplies the switching signalsto the first to fourth switching elements of the inverters.

Hereinafter, embodiments of the present invention are described withreference to the drawings.

First Embodiment

FIG. 1 shows the wireless power transmission device according to thefirst embodiment.

The wireless power transmission device is a wireless power device(power-transmission device) of the power-transmission side including apower-transmission unit 110, power-transmission unit 210, direct-current(DC) power supply 310, and drive device 312. The wireless powertransmission device wirelessly transmits power to the wireless powertransmission device (power-reception device) of the power-receptionside. Incidentally, although there are two power-transmission units inFIG. 1, a configuration including three or more power-transmission unitsmay be used as described later.

The DC power supply 310 is connected to both of the power-transmissionunit 110 and power-transmission unit 210 and supplies a DC power supplyto both of them as a drive source. Specifically, to one end in each ofthe power-transmission unit 110 and power-transmission unit 210, the DCpower supply 310 supplies a power-supply voltage (first power-supplyvoltage) and to the other end in each, the DC power supply 310 suppliesa ground voltage (second power-supply voltage).

The power-transmission unit 110 includes a single-phase full-bridgeinverter 120 and transmission coil unit 130. The single-phasefull-bridge inverter 120 is an inverter that operates as a DC-ACconverter, and includes switching elements 1201, 1202, 1203, and 1204and diodes (reflux diodes) 1201 a, 1202 a, 1203 a, and 1204 a connectedin inverse parallel to these switching elements 1201 to 1204. The“connected in inverse parallel” means that the flow direction of acurrent (electrical current) in each connected element is reverse (thedirection of current that reversely flows to the DC power supply). Theswitching elements 1201, 1202, 1203, and 1204 correspond to the first,the second, the third, and the fourth switching elements respectively,

One ends of the switching elements 1201 and 1202 are mutually connectedand one ends of the switching elements 1203 and 1204 are mutuallyconnected. The other ends of the switching elements 1201 and 1203 arecommonly connected to the power-supply terminal of the DC power supply310. In this way, power supply voltage is supplied from the DC powersupply 310. The other ends of the switching elements 1202 and 1204 arecommonly connected to a ground terminal of the DC power supply 310, thusground voltage is supplied from the DC power supply 310.

The connection node between the switching elements 1201 and 1202 isconnected to a terminal 1205 and a connection node between the switchingelements 1203 and 1204 is connected to a terminal 1206. A transmissioncoil unit 130 at least includes a coil 1301. One end of the transmissioncoil unit 130 (in this case, one end of the coil 1301) is connected to aterminal 1205 and the other end of the transmission coil unit 130 (inthis case, the other end of the coil 1301) is connected to the terminal1206. Here, the terminal 1205 corresponds to a positive output terminaland the terminal 1206 corresponds to a negative output terminal. Apotential difference between the terminal 1205 and terminal 1206corresponds to an output voltage of a single-phase full-bridge inverter120.

The single-phase full-bridge inverter 120, based on the power-supplyvoltage and ground voltage supplied from the DC power supply 310,generates AC power (AC voltage or AC current) by driving each switchingelement according to a switching signal supplied from a drive device312. When the switching element 1201 and switching element 1204 are on(“ON”) and the switching element 1202 and switching element 1203 are off(“OFF”), a current flows to the ground side of the DC power supply 310from the DC power supply 310 via the switching element 1201, coil 1301,and switching element 1204. When the switching element 1201 andswitching element 1204 are “OFF” and the switching element 1202 andswitching element 1203 are “ON”, the current flows to the ground side ofthe DC power supply 310 from the DC power supply 310 via the switchingelement 1203, coil 1301, and switching element 1202. As above, bygenerating a current that changes its direction by controlling ON/OFFswitching of each switching element, AC power is generated.

The single-phase full-bridge inverter 120 supplies the generated ACpower to a transmission coil unit 130. More specifically, an outputvoltage applied between the terminals 1205 and 1206 and a currentdecided in accordance with an impedance of the transmission coil unit130 flow, and in the coil 1301 of the transmission coil unit 130, anelectromagnetic field that corresponds to the current is generated. Theelectromagnetic field combines with a coil on the wireless powertransmission device (power-reception device) of the power-reception sideand as a result, power is transmitted (see FIG. 14 described later).

Further, the power-transmission unit 210 also has the same configurationas the power-transmission unit 110. That is, the power-transmission unit210 includes a single-phase full-bridge inverter 220 and transmissioncoil unit 230. The single-phase full-bridge inverter 220 includesswitching elements 2201, 2202, 2203, and 2204 and diodes (reflux diodes)2201 a, 2202 a, 2203 a, and 2204 a connected in inverse parallel tothese switching elements. The switching elements 2201, 2202, 2203, and2204 correspond to the first, the second, the third, and the fourthswitching elements respectively.

One ends of the switching elements 2201 and 2202 are mutually connectedand one ends of the switching elements 2203 and 2204 are mutuallyconnected. The other ends of the switching elements 2201 and 2203 arecommonly connected to the power-supply terminal of the DC power supply310. In this way, power supply voltage is supplied from the DC powersupply 310. The other ends of the switching elements 2202 and 2204 arecommonly connected to a ground terminal of the DC power supply 310, thusground voltage is supplied from the DC power supply 310.

The connection node between the switching elements 2201 and 2202 isconnected to a terminal 2205 and a connection node between the switchingelements 2203 and 2204 is connected to a terminal 2206. A transmissioncoil unit 230 at least includes a coil 2301. One end of the coil 2301 isconnected to a terminal 2205 and the other end of the coil 2301 isconnected to the terminal 2206.

The single-phase full-bridge inverter 220 generates AC power, based onthe power-supply voltage and ground voltage, by driving each switchingelement according to a switching signal supplied from a drive device312. Then, the generated AC power is supplied to the transmission coilunit 230. The coil 2301, upon receipt of the AC power from thesingle-phase full-bridge inverter 220, transmits power by magneticcoupling by combining with the coil on the wireless power transmissiondevice (power-reception device) side of the power-reception side.

The drive device 312 includes a switching signal generation circuit 311and drives a power-transmission unit 110 and power-transmission unit210. The switching signal generation circuit 311 generates a switchingsignal for driving the switching elements 1201 to 1204 of thesingle-phase full-bridge inverter 120 and a switching signal for drivingthe switching elements 2201 to 2204 of the single-phase full-bridgeinverter 220. Then, the switching signal generation circuit 311 suppliesthe generated switching signals to each switching element. Theseswitching signals are pulse waveform signals (see such as FIG. 4described later), and they have substantially the same duty ratio andfrequency. Hereafter, switching signals supplied to the switchingelements 1201 to 1204 and 2201 to 2204 may be expressed using the samereference numbers as the switching signals 1201 to 1204 and 2201 to2204.

The switching signal generation circuit 311 generates the switchingsignals 1201 to 1204 in order to complimentarily drive the switchingelement 1201 and switching element 1202, and switching element 1203 andswitching element 1204 in the single-phase full-bridge inverter 120.Also, the switching signal generation circuit 311 generates theswitching signals 2201 to 2204 in order to complimentarily drive theswitching element 2201 and switching element 2202, and switching element2203 and switching element 2204 in the single-phase full-bridge inverter220. In this way, AC power is generated in each single-phase full-bridgeinverter.

Here, by adjusting the phase relation of the switching signals 1201 and1203 in the single-phase full-bridge inverter 120, the switching signalgeneration circuit 311 is capable of adjusting the amplitude of theoutput voltage to the coil 1301. Similarly, by adjusting the phaserelation of the switching signals 2201 and 2203 in the single-phasefull-bridge inverter 220, the amplitude of the output voltage to thecoil 2301 can be adjusted. By adjusting the amplitude of the outputvoltage to the coils 1301 and 2301, the amplitude of the output currentto the coils 1301 and 2301 can be also adjusted. In addition, byadjusting the phase relation between the switching signal 1201 of thesingle-phase full-bridge inverter 120 and the switching signal 2201 ofthe single-phase full-bridge inverter 220, phase difference of theoutput voltage to the coils 1301 and 2301 are adjusted, and thus thephase difference of the output current to the coils 1301 and 2301 can beadjusted to a desired phase difference.

In the present embodiment, one of its characteristics is to reduceleakage of electromagnetic waves from the power transmission device toits surroundings using these functions to adjust amplitude of the outputvoltage and phase difference of the output voltage. In other words, apart of the electromagnetic field generated from the transmission coilunits 130 and 230 is emitted to their surroundings and becomes a leakedelectromagnetic field. The leaked electromagnetic field may affectperipheral devices of the power transmission device. Further, when thereare metals around, heat may be generated due to the leakedelectromagnetic field. For these reasons, the electromagnetic field thatleaks to the surroundings should preferably be kept small. In order toachieve this purpose, to mutually cancel out the electromagnetic fieldthat leaks from the transmission coil unit 130 and the electromagneticfield that leaks from the transmission coil unit 230, the amplitude andphase difference of the current in each transmission coil unit arecontrolled by adjusting the amplitude of the output voltage to eachtransmission coil unit and the phase difference of the output voltage.

When the transmission coil unit 130 and transmission coil unit 230include a coil of the similar characteristics, by controlling eachswitching signal so as to make an output voltage of the single-phasefull-bridge inverter 120 and single-phase full-bridge inverter 220 thesame amplitude and to become the opposite phase (180 degrees) (or tomake close to the opposite phase), their output currents become the sameamplitude and also the opposite phases. Subsequently, it is consideredthat the leaked electromagnetic fields become the same amplitude andalso opposite in the phases and thus the leaked electromagnetic fieldsare mutually cancelled. In such a case, at any point having the samedistance from where the two transmission coil units are arranged and ata point far enough from the transmission coil unit relative to its size,it is expected that the leaked electromagnetic fields are mutuallycancelled and become zero.

However, in reality, there may be variations in characteristics of thetransmission coil unit 130 and transmission coil unit 230 anddifferences in the connection state to the power-reception side. In suchcases, the impedance values of the transmission coil unit 130 andtransmission coil unit 230 become different, and even in a case whereoutput voltages of the single-phase full-bridge inverters 120 and 220become the same amplitude and also opposite phases, the current suppliedto the two transmission coil units and the generating electromagneticfield do not become the same. Accordingly, sufficient effect in theleaked electromagnetic field reduction cannot be expected. For thisreason, in order to supply the current having the same amplitude even insuch cases, output amplitudes of the two single-phase full-bridgeinverters are adjusted individually. Further, when the difference in thephase components of the impedance between the transmission coil unitscannot be ignored, the current phase difference does not become anopposite phase even when the phase difference of the output voltage ismade to an opposite phase. Accordingly, by adjusting the phase relationof the output voltage between the single-phase full-bridge inverters soas to make the phase difference of the current to become an oppositephase, the phase difference of the output current is made to an oppositephase. In this way, reduction of leaked electromagnetic field can beachieved sufficiently even when impedance of the transmission coil unitor the difference in the phase components cannot be ignored. Details ofsuch control of the switching signals for achieving leakedelectromagnetic field are described later.

Now, the relationship between the phase difference and attenuationamount is described. FIG. 2 shows the amplitudes of the leakedelectromagnetic field at a point where the distance from the twotransmission coil units are the same and the relationship of the phasedifference between currents of the two transmission coil units. Here,the amount of the current that flows through the two transmission coilunits is considered to be the same and normalized to the values at 0degrees. When the phase difference is 180 degrees (the opposite phase),theoretically the leaked electromagnetic field becomes “0”. Even whenthe phase difference is in the range of 180 degrees to +/−30 degrees,the leaked electromagnetic field is −10 dB or less, which is 1/10 orless and exhibits excellent cancelling effect. Accordingly, in thefollowing explanation, there may be used an expression of “adjusting thephase difference to 180 degrees”, however, it means to approach thephase difference to 180 degrees up to a degree where sufficientcancelling effect can be obtained. Specifically, it means to adjust thephase difference within the range of +/−30 degrees to make the phasedifference approach to 180 degrees. In addition, adjusting the phasedifference to any phase difference X other than 180 degrees means tomake the phase difference approach to a target value (phase differenceX) within a range of about X degrees +/−X/6 degrees. Further, when acancelling effect larger than 10 dB is required, the phase differencemay be approached to the target value in a narrower range.

Furthermore, in the present embodiment there are two power-transmissionunits. However, as described in other embodiments, the present inventioncan be more generally extended to a case where N number (an integer of 2or greater) of power-transmission units are included (see FIG. 15). Insuch cases, in order to achieve reduction in leaked electromagneticwaves, as to the phase difference between currents of thepower-transmission units, the phase difference between a current of the“M”th power-transmission unit and a current of the “M−1”thpower-transmission unit should be adjusted to 360×L/N degrees. “M” is aninteger of 2 or greater and “N” or smaller and “L” is an integer of 1 orgreater and smaller than “N”.

A specific example of the switching elements 1201 to 1204, and 2201 to2204 in FIG. 1 include semiconductor element such as FET and IGBT. FIG.1 shows the case of the FET element. Actions of the semiconductorelement are controlled by a signal supplied to a gate or base. Forexample, when using an N-type FET element, the semiconductor element is“ON” when the potential difference between the gate and the source is athreshold value or more, and “OFF” when the potential difference is lessthan the threshold value. Here, the switching signal supplied to theswitching element is the signal obtained by arranging a voltage signalthat makes the potential difference between the gate and the source to athreshold value or more and a voltage signal that makes the potentialdifference less than the threshold value at a predetermined duty ratioand frequency.

The following describes the case where the switching element becomes ONwhen the switching signal is at high level and the switching elementbecomes “OFF” when the switching signal is at low level. However, thesemay be inversed.

In the following, the reflux diodes 1201 a to 1204 a, and 2201 a to 2204a connected to the switching elements in the single-phase full-bridgeinverters 120 and 220 are described. One role of the reflux diode is,when changing the direction of the current flowing through the coil inON/OFF switching of each switching element (that is to inverse directionof the voltage applied to the coil), to protect each switching element.When switching ON/OFF of each switching element, due to inductance ofthe coil, direction of the current of the coil cannot be immediatelyinversed and thus a current that is inverse to the voltage applied tothe coil after switching flows. When using such as an IGBT, a switchingelement to which large current cannot be inversely flowed, by flowingthe current to the reflux diode connected in inverse parallel to theswitching element, an inverse current flows to each switching elementand thus prevents occurrence of damage and destruction of the element.Also, when using the switching signal including dead time describedlater, a period in which all the switching elements become “OFF” isgenerated during switching of the switching elements. In the case, byflowing the coil current to the reflux diode, occurrence of damage anddestruction of the switching element can be prevented. Incidentally, theconnecting position of the reflux diode is not limited to the positionsin FIG. 1 and may be changed according to the type of switching elementto be used The reflux diode is not essential in the point to exhibitfunctions of the present embodiment. Accordingly, as shown in FIG. 3, aconfiguration not connecting the reflux diode may be used.

The following describes, using FIG. 4, a relationship between theswitching signals 1201 to 1204 supplied to the switching elements 1201to 1204 of the single-phase full-bridge inverter 120 and the waveform ofthe output voltage from the single-phase full-bridge inverter 120.

FIG. 4(A), FIG. 4(B), and FIG. 4(C) show relationships of the switchingsignal and output voltage waveforms for the single-phase full-bridgeinverter 120. In the following description, voltage V_(IN) input fromthe DC power supply 310 is constant. A frequency for transmission is f₀[Hz]. In other words, the cycle of the transmission frequency is t₀=1/f₀[sec.]. The switching signals supplied to the four switching elementsall have the same duty ratio and are the pulse signals having the samecycle t₀. In FIG. 4(A) to FIG. 4(C), the duty ratio for the switchingsignals 1201 to 1204 is 50%.

FIG. 4(A) is an example where the switching signals 1201 to 1204 are setso that a fundamental wave component of the output voltage of thesingle-phase full-bridge inverter 120 becomes the maximum. In FIG. 4(A),the switching signal 1202 has a phase difference (phase lead) of 180degrees relative to the switching signal 1201.

Now, the phase difference is described. The phase difference means atime difference of the waveform between periodic signals. Phase lead of“P” degrees is equivalent to a time leading of P/360×t₀ for the periodicsignal having a cycle “t₀”. Also, in the case of periodic waveform, when360-degrees phase shift is performed, the waveform becomes the same. Forthis reason, phase lead of “P” degrees is equivalent to phase delay of“360−P” degrees. In the following explanation, expressions of “phaselead” and “phase delay” are used to explain the waveforms forconvenience. However, converting from phase delay to phase lead, andfrom phase lead to phase delay by phase shifting by 360 degrees areequivalent. Further, in the description, waveforms may be expressed innegative phase lead or negative phase delay. However, by using anabsolute value, they can be considered as expressing the waveformsequivalent to the waveforms expressed as positive phase delay or lead.In other words, to any “P”, phase delay of “P” degrees is defined to beequivalent to the phase lead of “−P” degrees, and the phase lead of “P”degrees is defined to be equivalent to the phase delay of “−P” degrees.

The switching signal 1203 has 180-degrees phase lead (T1 in FIG. 4(A))to the switching signal 1201. The switching signal 1204 has 180-degreesphase lead to the switching signal 1203. In other words, although theswitching signal 1204 has 360-degrees phase lead to the switching signal1201, the 360-degrees phase lead indicates a shift of time waveform forthe amount of 1 cycle, and thus equivalent to the 0-degrees phase leadwith respect to the periodic signal. Accordingly, the switching signal1201 is the same waveform as the switching signal 1204.

When the switching signals 1201 to 1204 are in such phase relations, asshown in FIG. 4(A), the output voltage of the single-phase full-bridgeinverter 120 becomes a rectangular waveform. In the figure, thebroken-line waveform shown overlapping to the rectangular waveformrepresents a fundamental wave component of the output voltage. Thefundamental wave component can be defined as the component of thefrequency f₀ for the waveform of the output voltage. The fundamentalwave component can be obtained by performing frequency resolution of thesignal of the output voltage and extracting only the fundamental wavecomponent.

FIG. 4(B) is, when compared to FIG. 4(A), an example for setting theamplitude of the fundamental wave component of the output voltage low.As in FIG. 4(A), the switching signal 1202 has 180-degrees phase lead tothe switching signal 1201. The switching signal 1203 has “180−P₁”degrees phase lead (T2 in FIG. 4(B)) to the switching signal 1201. P₁ isset to P₁>0. The switching signal 1204 has 180-degrees phase lead to theswitching signal 1203. Here, to the output voltage waveform, a period inwhich the output voltage becomes “0” (T3 in FIG. 4(B)) is inserted onlyby “t₀×P₁/360” seconds per 1 cycle as shown in FIG. 4(B). In this way,the amplitude of the fundamental wave component of the output voltagebecomes small compared to the case shown in FIG. 4(A).

FIG. 4(C) is, like FIG. 4(B), is another example for setting theamplitude of the fundamental wave component of the output voltage lowcompared to FIG. 4(A). In FIG. 4(B), a positive value was used for P₁.However, FIG. 4(C) is the case where a negative value is used. Theswitching signal 1203 has “180−P₁”-degrees phase lead (T4 in FIG. 4(C))to the switching signal 1201. P₁ is set to P₁<0. Also in the case, aperiod in which the output voltage becomes “0” (T5 in FIG. 4(C)) isinserted to the output voltage waveform only by “t₀×|P₁|/360” secondsper 1 cycle. In this way, the amplitude of the fundamental wavecomponent of the output voltage becomes small compared to FIG. 4(A).

As above, the magnitude of the fundamental wave component of the outputvoltage is determined by P₁. Any P₁ can be expressed within the range of−180 degrees to 180 degrees. To P₁ that is out of this range, by phaseshifting by 360 degrees times an integer. P₁ can be changed to becomewithin the range. When P₁ is expressed within the range of −180 degreesto 180 degrees, as |P₁| becomes small, the fundamental wave component ofthe output voltage becomes large, and as |P₁| becomes large, thefundamental wave component of the output voltage becomes small. In otherwords, by adjusting the value of P₁, the amplitude of the fundamentalwave component for the output voltage can be adjusted. FIG. 4(A)corresponds to the case where P₁ is “0” degrees. Incidentally, when P₁is −180 degrees or 180 degrees, ideally, the output voltage alwaysbecomes zero.

In the following, P₁ is expressed within the range of −180 degrees to180 degrees, and P₁ is called as an “amplitude adjustment parameter”.

In the above explanation, typically, an example of output amplitudeadjustment to the fundamental wave which is a frequency component havingthe maximum amplitude in the output voltage is shown. However, output ofthe single-phase full-bridge inverter includes harmonic wave componentof the fundamental wave in addition to the fundamental wave component.Also to the harmonic wave component, control of amplitude is possible bysimilarly adjusting |P₁|.

In FIG. 4(A) to FIG. 4(C), a case where a duty ratio of the switchingsignal is 50% is shown, In such case, when variations in the timing forthe switching signals are generated, the switching element 1201 andswitching element 1202 may simultaneously become “ON” and the switchingelement 1203 and switching element 1204 may simultaneously become “ON”.In such cases, output of the DC power supply 310 is short-circuited anda large current may be generated. Accordingly, by making the duty ratioless than 50%, a method of more safely preventing the switching elementsfrom becoming simultaneously “ON” is shown.

In FIG. 5, an example of the waveforms of each switching signal andoutput voltage when the duty ratio is less than 50% is shown. By makingthe duty ratio less than 50%, between the switching signal 1201 andswitching signal 1202, and between the switching signal 1203 andswitching signal 1204, the dead time DT where both of their levelsbecome a low level can be set. In this way, even when variations in thetiming for the switching signals are generated, turning the switchingelements simultaneously “ON” can be prevented. As above, even for thecase where the dead time DT is set, by setting the phase difference P₁which becomes a longer time difference than the dead time, like in FIG.4, amplitude of the fundamental wave component of the output voltage canbe controlled.

In the above, using the single-phase full-bridge inverter 120 as anexample, the relationships between the switching signals 1201 to 1204and output voltage waveform are explained. Also for the single-phasefull-bridge inverter 220, like the single-phase full-bridge inverter120, amplitude of the output voltage can be controlled. Also to thesingle-phase full-bridge inverter 220, like P₁ in the single-phasefull-bridge inverter 120, amplitude adjustment parameter can be definedand the parameter is shown as P₂. In other words, the output amplitudeof the single-phase full-bridge inverter 220 can be controlled with P₂using |P₂|.

From the above, the output amplitudes of the single-phase full-bridgeinverters 120 and 220 can be individually controlled by adjusting |P₁|and |P₂| for each. Subsequently, even when impedance of the transmissioncoil unit 130 and 230 differ, the output current of the same amplitudecan be generated, and thus leaked electromagnetic field having the sameamplitude can be generated.

On the other hand, as described above, in order to cancel out the leakedmagnetic field, the phase of the leaked electromagnetic field outputfrom each single-phase full bridge inverter needs to be inverted to anopposite phase (180 degrees). That is, to each transmission coil unit, acurrent of an opposite phase needs to be flowed. Between thetransmission coil units, when the difference of phase components ofimpedance is small enough to be ignored, the two single-phasefull-bridge inverters only need to be driven in an opposite phase (makethe phase difference of the output voltage 180 degrees between the twosingle-phase full-bridge Inverters). In this case, as the phasedifference of the current also becomes an opposite phase, the leakedelectromagnetic field also becomes an opposite phase.

In FIG. 6, examples of the switching signal and output voltage waveformswhen adjusting the phase difference of the output voltage to 180 degreesbetween two single-phase full-bridge inverter are shown.

An amplitude adjustment parameter for the single-phase full-bridgeinverter 120 is given as P₁, and an amplitude adjustment parameter forthe single-phase full-bridge inverter 220 is given as P₂. In FIG. 6,P₁>0, P₂>0, P₁<P₂. However, waveform can be similarly defined to any P₁and P₂ combinations.

In each pair of the switching signals 1201 and 1202, 1203 and 1204, 2201and 2202, and 2203 and 2204, there are 180-degrees phase differences.The switching signal 1203 has phase lead (T11) of “180−P₁” degrees tothe switching signal 1201, and the switching signal 2203 has phase lead(T12) of “180−P₂” degrees to the switching signal 2201.

Further, the switching signal 2201 has phase lead (T13) of“180−0.5(P₁−P₂)” degrees to the switching signal 1201. In the figure,T17 represents the period of difference between the phase lead“1.80−0.5(P₁−P₂)” degrees and 180 degrees and its length is“0.5×t₀×|P₁−P₂|/360” seconds.

To any P₁ and P₂, by providing the switching signals set for such phaserelation, a desired amplitude can be obtained while using thefundamental wave component of the output voltage of the two single-phasefull-bridge inverter as an opposite phase (T14). Incidentally, T15represents a period where the output voltage of the single-phasefull-bridge inverter 120 becomes “0”, and its length is “t₀×|P₁|/360”seconds. T16 represents a period where the output voltage of thesingle-phase full-bridge inverter 220 becomes “0”, and its length is“t₀×|P₂|/360” seconds.

When difference of the phase components of impedance betweentransmission coil units cannot be ignored at this point, even when thephase difference of the output voltage is made to an opposite phase, thedifference of the current phase does not become an opposite phase sothat the phase difference of the leaked electromagnetic field does notbecome an opposite phase and the effect in leaked magnetic fieldreduction reduces. For this reason, to make the phase difference of thecurrent an opposite phase, appropriately setting the phase difference ofthe output voltage between the single-phase full-bridge inverters andmaking the phase of the leaked electromagnetic field an opposite phaseare preferable.

For the case, the switching signal 2201 should be adjusted to have phaselead of “PP₁−0.5(P₁−P₂)” degrees relative to the switching signal 1201.For other switching signals 2202 to 2204 and 1202 to 1204, the samerelationship should be maintained relative to the switching signal 2201and switching signal 1201 respectively. Here, PP₁ represents the currentphases of the two transmission coil units, in other words, the phasedifference between the output voltages of the single-phase full-bridgeinverter 120 and single-phase full-bridge Inverter 220 where the phasesof the leaked electromagnetic fields generated by the two transmissioncoil units become opposite phases, “PP₁” is called “a current phaseadjustment parameter”.

In FIG. 7, an example of the waveforms of the switching signal andoutput voltage when the phase difference of the output voltages is setto PP₁ degrees between the two single-phase full-bridge inverters. Theexample shown in FIG. 6 before corresponds to the case where PP₁=180degrees.

In FIG. 7, the switching signal 1203 has the phase lead (T21) of“180−P₁” degrees relative to the switching signal 1201. The switchingsignal 2203 has the phase lead (T22) of “180−P₂” degrees relative to theswitching signal 2201.

Further, the switching signal 2201 has phase lead (T23) of“PP₁−0.5(P₁−P₂)” degrees relative to the switching signal 1201. To anyP₁ and P₂, by providing a switching signal set to such phase relation, adesired amplitude can be obtained while making the phase differencebetween the fundamental wave components of the output voltages of thetwo single-phase full-bridge inverters to PP₁ degrees (T24). Inaddition, T25 represents a period where the output voltage of thesingle-phase full-bridge inverter 120 becomes “0” and its length is“t₀×|P₁|/360” seconds. T26 represents a period where the output voltageof the single-phase full-bridge inverter 220 becomes “0” and its lengthis “t₀×|P₂|/360” seconds. T27 represents a period of difference betweenthe period of T23 and 180 degrees, and its length ist₀×|PP₁−180+0.5|P₁−P₂|/360 seconds.

Now, cases where it becomes effective to adjust the phase difference ofthe output voltage to 180 degrees and to degrees other than 180 degreesare specifically described. Also, when adjusting to degrees other than180 degrees, specifically to what value the phase difference of theoutput voltage should be set is described.

In FIG. 8(A), a case where the phase lead Q₁ of the output currentrelative to the output voltage of the single-phase full-bridge inverter120 and the phase lead Q₂ of the output current relative to the outputvoltage of the single-phase full-bridge inverter 220 are substantiallythe same. The solid line represents a fundamental wave component of thevoltage and a broken line represents a fundamental wave component of thecurrent. In this case, by adjusting the phase difference of thefundamental wave component of the voltage of the single-phasefull-bridge inverter 220 relative to the fundamental wave component ofthe voltage of the single-phase full-bridge inverter 120 to 180 degrees(T31), the current phase difference also becomes approximately180-degrees phase difference (T32). Accordingly, effective effect ofleaked electromagnetic field can be expected.

In FIG. 8(B), a case where the phase lead Q₁ of the output currentrelative to the output voltage of the single-phase full-bridge inverter120 and the phase lead Q₂ of the output current relative to the outputvoltage of the single-phase full-bridge inverter 220 are different isshown. When Q₁ and Q₂ are different, by setting the phase difference ofthe fundamental wave component of the voltage of the single-phasefull-bridge inverter 220 relative to the fundamental wave component ofthe voltage of the single-phase full-bridge inverter 120 to the phasedifference of other than 180 degrees, that is the phase lead (T33) of180+Q₁−Q₂ degrees, the current waveform becomes an opposite phase (T34)and a great effect in the leaked electromagnetic field reduction can beexpected. Further, in FIG. 8(B), Q₁<0 and Q₂>0 and includes the waveformwhen the phase lead Q₁ is negative. However, as defined in the above, itis equivalent to the phase delay of Q₁ degrees. Also, similarexplanation can be made for examples with Q₁ and Q₂ of any sign and anyvalue other than the one shown in FIG. 8(B).

Further, a coil that corresponds to either of the two single-phasefull-bridge inverters may be arranged so as to generate anelectromagnetic field having an opposite direction relative to theoutput current direction from the single-phase full-bridge inverter. Insuch a case, as a great cancelling effect can be obtained for thecurrent of the phase difference at 0 degrees between the single-phasefull-bridge inverters, PP₁ should be adjusted to 0 degrees or around(that is a value taking into account Q₁ and Q₂). Such arrangement ispossible by, in a case of a spiral-type or solenoidal-type coil,changing the direction of the winding wire to the opposite direction.

FIG. 9 shows a configuration example of a switching signal generationcircuit 311. The switching signal generation circuit 311 includes phaseshifters 201 to 207. The switching signals 1201 to 1204 and 2201 to 2204are generated from the reference signal using the phase shifters. Thereference signal 200 is a pulse signal having the same duty ratio andthe same frequency as those of other switching signals. The referencesignal 200 can be generated by using such as phase-locked loop (PPL).The switching signal generation circuit 311 may include a signalgeneration circuit that generates the reference signal 200.

The switching signal 1201 uses the reference signal 200 as it is. Theswitching signal 1202 has the phase lead of 180 degrees relative to theswitching signal 1201 so that it can be generated by providing phaselead (phase difference) of 180 degrees to the reference signal by thephase shifter 201. The switching signal 1203 has the phase lead of“180−P₁” degrees relative to the switching signal 1201 so that byproviding the phase lead by the phase shifter 202 the switching signal1203 can be generated. The switching signal 1204 has the phase lead of180 degrees relative to the switching signal 1203 so that by providingthe phase lead of 180 degrees to the switching signal 1203 by the phaseshifter 203, the switching signal 1204 can be generated.

The switching signal 2201 has the phase lead of (PP₁−0.5(P₁−P₂)) degreesrelative to the switching signal 1201 so that it can be generated byproviding the phase lead to the switching signal 1201 by the phaseshifter 204. The switching signal 2202 has the phase lead of 180 degreesrelative to the switching signal 2201 so that it can be generated byproviding the phase lead of 180 degrees to the switching signal 2201 bythe phase shifter 205. The switching signal 2203 has the phase lead of“180−P₂” degrees relative to the switching signal 2201 so that it can begenerated by providing the phase lead to the switching signal 2201 bythe phase shifter 206. The switching signal 2204 has the phase lead of180 degrees relative to the switching signal 2203 so that it can begenerated by providing the phase lead of 180 degrees to the switchingsignal 203 by the phase shifter 207.

Here, “providing the phase lead of “R” degrees” means the same asproviding the phase delay of “360−R” degrees. For this reason, eachphase shifter can be configured in either way to provide phase delay orphase lead.

Further, the phase shifter may include a delay device providing thesimilar effect. The phase shifter providing the phase lead of “R”degrees is equivalent to the phase delay of “360−R” degrees.Accordingly, the phase shifter providing the phase lead of “R” degreescan be replaced by a delay device which generates a delay oft₀×(360−R)/360 seconds to the cycle t₀.

Further, in a case of a waveform having 50% duty, the 180-degrees phaseshifter may be configured from an inverter that reverses between a highlevel and low level.

FIG. 10 shows another configuration example of the switching signalgeneration circuit 311. The switching signal generation circuit 311includes the phase shifters 211 to 217. The switching signal 1201 usesthe reference signal 200 as it is. The switching signal 1202 isgenerated by providing 180-degrees phase lead to the reference signal200 by the phase shifter 211. Other switching signals 1203, 1204, and2201 to 2204 can be generated by providing phase lead for thecorresponding phase lead amount to these two switching signals 1201 and1202 by the phase shifter 212 to 217. In other words, the switchingsignals 1203 and 1204 are generated by providing 180-degrees phase leadto the switching signals 1201 and 1202 by the phase shifter 212 and 213.The switching signal 2201 and 2202 are generated by providing the phaselead of “PP₁−0.5(P₁−P₂)” degrees to the switching signals 1201 and 1202by the phase shifters 214 and 215. The switching signals 2203 and 2204are generated by providing the phase lead of “180+PP₁−0.5(P₁+P₂)”degrees to the switching signals 1201 and 1202 by the phase shifter 216and 217.

FIG. 11 further shows another configuration example of the switchingsignal generation circuit 311. The switching signal generation circuit311 includes the phase shifters 221 to 227. The switching signal 1201uses the reference signal 200 as it is. Other switching signals 1202 to1204, and 2201 to 2204 are generated by providing corresponding phaselead to the switching signal 1201 by each of the phase shifters 221 to227. In other words, the switching signal 1202 is generated by providing180-degrees phase lead to the switching signal 1201 by the phase shifter221. The switching signal 1203 is generated by providing the phase leadof “180−P₁” degrees to the switching signal 1201 by the phase shifter222. The switching signal 1204 is generated by providing the phase leadof “−P₁” degrees to the switching signal 1201 by the phase shifter 223.The switching signal 2201 is generated by providing the phase lead of“PP₁−0.5(P₁−P₂)” degrees to the switching signal 1201 by the phaseshifter 224. The switching signal 2202 is generated by providing thephase lead of “180+PP₁−0.5(P₁−P₂)” degrees to the switching signal 1201by the phase shifter 225. The switching signal 2203 is generated byproviding the phase lead of “180+PP₁−0.5(P₁−P₂)” degrees to theswitching signal 1201 by the phase shifter 226. The switching signal2204 is generated by providing the phase lead of “PP₁−0.5(P₁+P₂)”degrees to the switching signal 1201 by the phase shifter 227.

Any other configuration generating a switching signal can be used aslong as the switching signal combinations that satisfy a predeterminedphase relation are obtained.

The transmission coil unit 130 and transmission coil unit 230 shown inFIG. 1 include the coil 1301 and coil 2301 respectively, however, otherelements may be included to the transmission unit. FIG. 12(A) and FIG.12(B) show other configurations of the transmission coil unit,

In FIG. 12(A), the capacitive element 332 is serially connected to thecoil 331 and accordingly, a resonator for LC is formed. In the presentexample, the capacitive element 332 is connected to the positive outputterminal 333, however, the capacitive element may be connected to thenegative output terminal 334. In FIG. 12(B), a coil 341 and a capacitiveelement 342 are connected in parallel between the positive outputterminal 343 and negative output terminal 344. In this way, a resonatorfor LC is formed. By combining configurations of FIG. 12(A) and FIG.12(B), the capacitive element may be connected both in serial andparallel relative to the coil.

Further, in FIG. 1, one DC power supply 310 is commonly connected to thesingle-phase full-bridge inverters 120 and 220. However, the DC powersupply may be connected separately for each single-phase full-bridgeinverter.

FIG. 13 shows a configuration example where DC power supply is connectedseparately for each single-phase full-bridge inverter. To thesingle-phase full-bridge inverter 120, a DC power supply 410, and to thesingle-phase full-bridge inverter 220, a DC power supply 510 isconnected. In these cases, a part of the amplitude control for theoutput voltage of the single-phase full-bridge inverters 120 and 220 maybe performed by amplitude adjustment of the output DC voltage of the DCpower supply 410 and DC power supply 510, and a part of the remainingamplitude control by adjustment using the amplitude adjustmentparameters P₁ and P₂ previously described. For example, in amplitudecontrol of the output voltages of the DC power supplies 410 and 510,rough adjustment of amplitude may be performed by large variation unit(step width) and finer adjustment may be performed by adjustment usingthe amplitude adjustment parameters P₁ and P₂.

FIG. 14 shows the power transmission device (a wireless powertransmission device on the power-transmission side) illustrated in FIG.1, and wireless power transmission device Including the power-receptiondevice (wireless power transmission device on the power-reception side).The power-reception device includes two power reception coil units 610and 710, rectifiers (AC/DC converters) 810 and 910, and a load 1010. Thepower reception coil units 610 and 710 include one coil, coil 6101 andcoil 7101 respectively.

Through combining of the transmission coil unit 130 with power receptioncoil unit 610 and combining of the transmission coil unit 230 with powerreception coil unit 710, power is wirelessly transmitted. Thetransmitted power is converted into direct current by the rectifiers 810and 910, and supplied to the load 1010. The load 1010 is a device whichconsumes or stores the supplied direct-current power.

The configuration of the power-reception device is not limited to theconfiguration shown in FIG. 14. For example, between the rectifier andthe load 1010, a DC-DC converter may be further included. In addition,the load may be connected separately for each power reception coil unitinstead of connecting commonly to the power reception coil units 610 and710. The power reception coil unit is not limited to include only onecoil, and various changes are possible as shown in FIG. 12 as in thetransmission coil unit. Further, in the configuration shown in FIG. 14,the power transmitted from the two coils are received by correspondingtwo coils, however, there may be 1 or 3 or more coils on thepower-reception side. For example, the configuration where the powertransmitted from the two coils is received by one coil or 3 or morecoils may be used.

As above, according to the present embodiment, by adjusting the phase ofthe switching signal driving the single-phase full-bridge inverterarranged corresponding to the transmission coil unit, voltage amplitudeand voltage phase are individually adjusted for each transmission coilunit. In this way, even when characteristics and arrangement, etc. ofthe transmission coil units are not symmetrical, phase differencebetween the currents of the transmission coil units can be made to aparticular relation (reversed phase), and also the amplitude of thecurrent in each transmission coil unit can be controlled so as tomutually cancel out their leaked electromagnetic fields.

Second Embodiment

FIG. 15 shows the wireless power transmission device according to thesecond embodiment. In the first embodiment shown in FIG. 1, there weretwo power-transmission units. However, in the present embodiment, thenumber of the power-transmission units are extended to 3. For elementshaving the same name as in FIG. 1, common symbols are assigned andexcept for those changed and extended processes, repeated explanationsare omitted.

In addition to the power-transmission units 110 and 210, apower-transmission unit 1110 is added. The power-transmission unit 1110includes a single-phase full-bridge inverter 1120 and a transmissioncoil unit 1130. The single-phase full-bridge inverter 1120 includes fourswitching elements 11201, 11202, 11203, and 11204, and diodes 11201 a,11202 a, 11203 a, and 11204 a connected in inverse parallel to theswitching elements respectively. Each of the switching elements 11201,11202, 11203, and 11204 corresponds to the first, the second, the third,and the fourth switching elements respectively. The transmission coilunit 1130 includes a coil 11301. The single-phase full-bridge inverter1120 and the transmission coil unit 1130 are connected via the terminals11205 and 11206. The configuration of the single-phase full-bridgeinverter 1120 is the same as the single-phase full-bridge inverter 120or 220 so that its explanation is omitted.

When three power-transmission units are used, in order to obtain aneffective effect in leaked electromagnetic field reduction, the currentphase of each coil should be set so as to vary by 360/3=120 degrees. Avector diagram of the leaked electromagnetic fields after adjusting thephase relation as above, and also after adjusting the amplitudes of theleaked electromagnetic fields to match is shown in FIG. 16. As shown, agreat reduction effect can be obtained even for the case of threephases.

To make the current phase of each transmission coil unit to vary by 120degrees, the phase of the fundamental wave component of the outputvoltage of the single-phase full-bridge inverter 220 should be set so asto lead 120 degrees relative to the single-phase full-bridge inverter120. This setting corresponds to setting PP₁=120 degrees in the firstembodiment. In addition, the phase of the fundamental wave component ofthe output voltage of the single-phase full-bridge inverter 1120 shouldbe set to lead 240 degrees relative to the single-phase full-bridgeinverter 120. When the phase lead in the fundamental wave component ofthe output voltage of the single-phase full-bridge inverter 1120relative to the single-phase full-bridge inverter 120 is expressed usingthe current phase adjustment parameter PP₂, this setting corresponds tosetting PP₂ to 240 degrees. Here, the phase lead Q₁ relative to thevoltage of the output current from the single-phase full-bridge inverter120, phase lead Q₂ of the output current relative to the output voltagefrom the single-phase full-bridge inverter 220, and phase lead Q₃ of theoutput current relative to the output voltage from the single-phasefull-bridge inverter 1120 are assumed to be substantially the same.

In FIG. 17, when PP₁ and PP₂ are set as above, an example of thewaveform of the switching signal the switching signal generation circuit311 provides to each single-phase full-bridge inverter is shown. Likethose explained above, when the amplitude adjustment parameters are P₁,P₂, and P₃ (P₁, P₂, and P₃>0), the switching signal 1203 has the phaselead (T41) of “180−P₁” degrees relative to the switching signal 1202,the switching signal 2203 has the phase lead (T42) of “180−P₂” degreesrelative to the switching signal 2201, and the switching signal 11203has the phase lead (143) of “180−P₃” degrees relative to the switchingsignal 11201. The switching signal 2201 has the phase lead (T44) of“120−0.5(P₁−P₂)” degrees relative to the switching signal 1201. P₁−P₂<0.Also, the switching signal 11201 has the phase lead (T45) of“240−0.5(P₁−P₃)” degrees relative to the switching signal 1101. P₁−P₃<0.By setting as above, the fundamental wave component of the outputvoltage from the single-phase full-bridge inverter 220 possesses120-degrees phase lead (T46) relative to the fundamental wave componentof the output voltage from the single-phase full-bridge inverter 120,and the fundamental wave component of the output voltage from thesingle-phase full-bridge inverter 1120 possesses 240-degrees phase lead(T47) relative to the fundamental wave component of the output voltagefrom the single-phase full-bridge inverter 120.

Here, the case where the phase lead Q₁, Q₂, and Q₃ of the output currentrelative to the output voltage from each single-phase full-bridgeinverter are assumed to be substantially the same. However, if theirphase leads differ, as described in the first embodiment, the phasedifferences of the current should be adjusted so as to vary by 120degrees taking into account the differences in these phase leads Q₁, Q₂,and Q₃. For example, the phase difference of the fundamental wavecomponent of the voltage of the single-phase full-bridge inverter 220relative to the fundamental wave component of the voltage of thesingle-phase full-bridge inverter 120 should be set to “120+Q₁−Q₂”degrees, and the phase difference of the fundamental wave component ofthe voltage of the single-phase full-bridge inverter 1120 relative tothe fundamental wave component of the voltage of the single-phasefull-bridge inverter 120 should be set to “240+Q₁−Q₃” degrees. Q₁<0,Q₂>0, Q₃>0. Here, adjustment of the phase difference (adjustment takinginto account the variation of the above phase leads) relative to thefundamental wave component of the voltage is performed using the firstsingle-phase full-bridge inverter 120 as the basis. However, thesingle-phase full-bridge inverter as the basis may be the secondsingle-phase full-bridge inverter 220 or another single-phasefull-bridge inverter.

In FIG. 15, a case with three power-transmission units are shown.However, in the present invention, more generally, the case can beextended to include “N” number (an integer of 2 or more) ofpower-transmission units. In such the case, the phase difference betweenthe fundamental wave components of the output currents from thesingle-phase full-bridge inverters in the “M”th and “M−1”thpower-transmission units should be adjusted to become “360×L/N” degrees.“M” is an integer of 2 or greater and N or less. “L” is an integer of 1or greater and less than “N”. In this way, the output current of theN-phase can be obtained and N-phase leaked electromagnetic fields aregenerated from the “N”th transmission coil units. However, by havingthese leaked electromagnetic fields canceled out each other, reductionof the leaked electromagnetic fields can be achieved. Further, when thephase leads relative to the voltage of the output current from eachsingle-phase full-bridge inverter are the same, by adjusting the phasedifference in the fundamental wave component of the output voltage tobecome “360×L/N” degrees, the phase difference in the fundamental wavecomponent of the output current is adjusted to “360×L/N” degrees.

FIG. 18 shows a configuration example of the switching signal generationcircuit 311 when “N” (integer of 2 or greater) number ofpower-transmission units are included. The configuration shown in FIG.18 is obtained by extending the configuration shown in FIG. 9 to the “N”phase. The phase difference between the fundamental wave components ofthe “M”th single-phase full-bridge inverter and the single-phasefull-bridge inverter 120 (the current phase adjustment parameterrelative to the “M”th single-phase full-bridge inverter) is representedas “PP_(M−1)”. Further, the amplitude adjustment parameter relative tothe “M”th single-phase full-bridge inverter is represented as “P_(M)”.Furthermore, like in FIG. 10 and FIG. 11, even for the case of N-phase,various configurations for generating the switching signal are possible.In the following, what differs from FIG. 9 is explained.

The switching signal 11201 relative to the switching element 11201 ofthe third single-phase full-bridge inverter 1120 (see FIG. 15) isgenerated by providing the phase lead of “PP₂−0.5(P₁−P₂)” degrees to thereference signal 200. The switching signal 11202 for the switchingelement 11202 is generated by providing 180-degrees phase lead to theswitching signal 11201 by the phase shifter 502. The switching signal11203 for the switching element 11203 is generated by providing thephase lead of “180−P₃” degrees to the switching signal 11201 by thephase shifter 503. The switching signal 11204 for the switching element11204 is generated by providing 180-degrees phase lead to the switchingsignal 11203 by the phase shifter 504.

Further, the switching signal M01 for the first switching element of the“M”th single-phase full-bridge inverter is generated by providing thephase lead of “PP_(M−1)−0.5(P₁−P_(M))” degrees to the reference signal200 by the phase shifter 511. The switching signal M02 for the secondswitching element is generated by providing the 180-degrees phase leadto the switching signal M01 by the phase shifter 512. The switchingsignal M03 for the third switching element is generated by providing thephase lead of “180−P_(M)” degrees to the switching signal M01 by thephase shifter 513. The switching signal M04 for the fourth switchingelement is generated by providing 180-degrees phase lead to theswitching signal M03 by the phase shifter 514.

As above, according to the present embodiment, even when the number ofthe power-transmission units is three or more, a similar effect as inthe first embodiment can be obtained by appropriately setting the phaserelation of the switching signal between respective single-phasefull-bridge inverters.

Third Embodiment

FIG. 19 shows the wireless power transmission device according to thethird embodiment,

FIG. 19 includes a current amplitude detection circuit 140 and a currentamplitude detection circuit 240 in addition to the configuration inFIG. 1. Also, in addition to the switching signal generation circuit311, the drive device 312 includes a parameter determination circuit 313where an amplitude adjustment parameter is calculated.

The current amplitude detection circuit 140 detects the amplitude of theoutput current of the single-phase full-bridge inverter 120 and notifiesthe detected information indicating the amplitude to the drive device312. The current amplitude detection circuit 240 detects the amplitudeof the output current of the single-phase full-bridge inverter 220 andnotifies the detected information indicating the amplitude to the drivedevice 312.

The parameter determination circuit 313 in the drive device 312 adjuststhe values of the amplitude adjustment parameters P₁ and P₂ so as tomake the amplitude differences in each output current small and forexample, to make the difference to approach “0” based on the informationobtained from the current amplitude detection circuits 140 and 240.

Specifically, when the amplitude of the output current from thesingle-phase full-bridge inverter 120 is larger than the output currentamplitude of the single-phase full-bridge inverter 220, P₁ is changed tomake |P₁| larger in order to decrease the output current of thesingle-phase full-bridge inverter 120. Or, to increase the outputcurrent of the single-phase full-bridge inverter 220, P₂ is changed soas to make |P₂| small. In this way, the difference in amplitude betweencurrents can be made smaller.

The switching signal generation circuit 311, in accordance with theparameters P₁ and P₂ calculated by the parameter determination circuit313, generates the switching signals 1201 to 1204 and 2201 to 2204 andsupplies the switching signals to the single-phase full-bridge inverters120 and 220. For the current phase difference adjustment parameter PP₁,a value provided in advance should be used. For example, when influencedue to the difference between Q₁ and Q₂ of the single-phase Full-bridgeinverters 120 and 220 can be ignored, PP₁ should be set to 180 degrees.

In FIG. 19, although the output current of each single-phase full-bridgeinverter was detected, similarly, an input current of each single-phasefull-bridge inverter may be detected and controlled to match theiramplitudes of the input current. In a typical configuration, there isenough correlation between an input current and output current, so thatan object can be achieved even under this configuration. Theconfiguration of the current amplitude detection circuits 140 and 240when detecting the input current is shown in broken line in the figure.

FIG. 20 shows another example of the wireless power transmission deviceaccording to the third embodiment. The current amplitude detectioncircuits in FIG. 19 are replaced with the current amplitude/phasedetection circuits 141 and 241. In addition, to the drive device 312, aparameter determination circuit 314 is added in addition to theparameter determination circuit 311

The current amplitude/phase detection circuit 141 detects the amplitudeand phase of the output current from the single-phase full-bridgeinverter 120 and notifies the detected information representing theamplitude and phase to the drive device 312. The current amplitude/phasedetection circuit 241 detects the amplitude and phase of the outputcurrent from the single-phase full-bridge inverter 220 and notifies thedetected information representing the amplitude and phase to the drivedevice 312.

The parameter determination circuit 313, like in FIG. 19, adjusts theamplitude adjustment parameters P₁ and P₂ to make the amplitudedifferences of each output current small. Whereas, the parameterdetermination circuit 314 adjusts the current phase differenceadjustment parameter PP₁ to make the detected phase difference approach180 degrees.

The switching signal generation circuit 311 generates the switchingsignals 1201 to 1204 and 2201 to 2204 in accordance with the parametersP₁ and P₂ calculated by the parameter determination circuits 313 and314, and supplies them to the single-phase full-bridge inverters 120 and220.

In FIG. 19 and FIG. 20, when configurations of the transmission coilunits 130 and 230 vary, the output current amplitude of eachsingle-phase full-bridge inverter may be adjusted. For example, when thenumber of windings differs between the coil 1301 and coil 2301, theintensity of the generating electromagnetic fields can be madesubstantially the same by adjusting the current amplitudes to make thecurrent amplitudes become the ratio that corresponds to the ratio of thenumber of windings.

Further, when the place where the leaked electromagnetic field is to bereduced and the positional relation of each transmission coil unit areknown, each current amplitude may be adjusted according to thepositional relation. For example, each current amplitude may be adjustedso as to become the ratio that corresponds to the ratio of the distancesbetween the place and each of the transmission coil units 1 and 2.

In FIG. 19 and FIG. 20, the case of using two power-transmission unitsare shown, however, embodiment using 3 or more is possible.

The current amplitude detection circuit 140 and current amplitudedetection circuit 240 shown in FIG. 19 and the current amplitude/phasedetection circuit 141 and current amplitude/phase detection circuit 241are disposed outside the drive device 312. However, these elements maybe disposed inside the drive device 312.

All or a part of the parameter determination circuits 313 and 314 andswitching signal generation circuit 311 shown in FIG. 19 and FIG. 20 maybe achieved with hardware such as a processor, EPGA, and ASIC. When aprocessor is used, functions of such processing sections can beperformed by the processor reading and executing the program stored inadvance in a recording medium such as a memory and SSD.

As above, according to the present embodiment, by controlling theswitching signals from each single-phase full-bridge inverter to makethe amplitude differences between respective output currents small bydetecting amplitude of the output current from each single-phasefull-bridge inverter, the leaked electromagnetic fields can be reduced.In addition, by detecting the amplitude and phase of the output currentfrom each single-phase full-bridge inverter to make the amplitudedifferences between respective output currents small, and by controllingthe phase of the switching signal so as to make the phase difference ofthe current 180 degrees, the leaked electromagnetic fields can bereduced even when impedance characteristics in each transmission coilunit differs,

Fourth Embodiment

FIG. 21 shows an example of the wireless power transmission deviceaccording to the fourth embodiment. The device in FIG. 21 detects aleaked electromagnetic field and in accordance with the intensity of theleaked magnetic field, adjusts the amplitude adjustment parameter andcurrent phase adjustment parameter. The device in FIG. 21 includes, inaddition to the configuration in FIG. 1, a leaked electromagnetic fielddetection circuit 316. Also, the drive device 312 includes a parameterdetermination circuit 315 in addition to the switching signal generationcircuit 311.

The leaked electromagnetic field detection circuit 316 detectselectromagnetic fields at predetermined places as the leakedelectromagnetic field of the wireless power transmission device, andnotifies the information representing the intensity of the detectedleaked electromagnetic field to the drive device 312. The places fordetecting the electromagnetic field may be inside the wireless powertransmission device or outside the wireless power transmission device.In the latter case, the leaked electromagnetic field detection circuit316 may be disposed apart from the wireless power transmission device.When doing so, by installing a wireless IF or wired IF to both of theleaked electromagnetic field detection circuit 316 and wireless powertransmission device, the information representing the intensity of thedetected electromagnetic field may be notified to the wireless powertransmission device from the leaked electromagnetic field detectioncircuit 316 in the wireless or wired manner. In this example, the leakedelectromagnetic field detection circuit 316 detected the electromagneticfield, however, a magnetic field, electric field, or both of these maybe detected instead. Detecting the magnetic field or electric field alsoenables to find out the intensity of the leaked electromagnetic field.

Based on the information obtained from the leaked electromagnetic fielddetection circuit 316, the parameter determination circuit 315 adjuststhe parameters P₁, P₂, and PP₁ so as to make the leaked electromagneticfield small. As for the change method, for example, one from theparameters P₁, P₂, and PP₁ is increased (or decreased) for apredetermined amount and the intensity of the leaked electromagneticfield before and after the change is compared. When the leakedelectromagnetic field is decreased, the parameter is increased (ordecreased) for the predetermined amount, and when the leakedelectromagnetic field is increased, the parameter is decreased (orincreased) for a predetermined amount. By repeating this change, eachparameter can be adjusted to make the leaked electromagnetic field assmall as possible. In contrast, by setting a threshold, the parameteradjustment may be repeated until the intensity of the leakedelectromagnetic field becomes the threshold or below.

At this point, the parameter determination circuit 315 may temporarilyset “0” to PP₁, that is, to set such that the leaked electromagneticfields generated from each transmission coil unit become in phase, andset “0” for PP₁ when the intensity of the detected leakedelectromagnetic field is the threshold or less. The threshold representsthe upper limit of the acceptable leaked electromagnetic fieldintensity. When performing transmission such that the leakedelectromagnetic fields become in phase, the electromagnetic fieldsgenerated from the two transmission coil units interfere to make themmutually stronger compared to the case where transmission is performedto make the leaked electromagnetic fields the opposite phases.Accordingly, an advantageous effect of transmitting a greater power witha small current can be expected,

FIG. 22 shows another example of, the wireless power transmission deviceaccording to the fourth embodiment. The leaked electromagnetic fielddetection circuit 316 in FIG. 21 is replaced with the leakedelectromagnetic field detection circuits 317 and 318. Although theleaked electromagnetic field detection circuit 316 is arranged to theplace where decrease of the leaked electromagnetic field is desired inFIG. 21, however, in FIG. 22, the leaked electromagnetic field detectioncircuits 317 and 318 are arranged at any place desired.

Each of the leaked electromagnetic field detection circuits 317 and 318detects the leaked electromagnetic field in the place they werearranged, and notify the information representing the intensity of thedetected leaked electromagnetic field to the drive device 312. Like inFIG. 19, the leaked electromagnetic field detection circuits 317 and 318may be arranged inside the wireless power transmission device or outsidethe wireless power transmission device.

The parameter determination circuit 319 in the drive device 312, basedon the information obtained from the leaked electromagnetic fielddetection circuits 317 and 318, adjusts the parameters P₁, P₂, and PP₁so as to make the ratio of the intensity of each leaked electromagneticfield to become the predetermined value or approach the predeterminedvalue. The predetermined value is determined based on the positionalrelation of the place where decrease of the leaked electromagnetic fieldis required and the locations of the leaked electromagnetic fielddetection circuits 317 and 318.

For example, when the leaked electromagnetic field detection circuits317 and 318 are arranged proximally to the transmission coil unit 130and transmission coil unit 230, adjust the parameter so as to make theratio of the outputs from these two detection circuits “1”. In this way,the amounts of the electromagnetic fields produced by these twotransmission coil units become substantially the same. In such the case,at a point having the same distance from the two transmission coil unitsand at a point having enough distance relative to the coil size, leakedelectromagnetic fields are mutually cancelled and a great reductioneffect can be expected.

Although a case of using two transmission coil units is shown in FIG. 21and FIG. 22, embodiments using three or more transmission coil units isalso possible.

All or a part of the parameter determination circuits 315 and 319 andswitching signal generation circuit 311 shown in FIG. 21 and FIG. 22 maybe achieved with hardware such as a processor, EPGA, and ASIC. When aprocessor is used, functions of such processing sections can beperformed by the processor reading and executing the program stored inadvance in a recording medium such as a memory and SSD.

As above, according to the present embodiment by detecting a leakedelectromagnetic field at a predetermined place and adjusting theamplitude adjustment parameters P₁ and P₂ and current phase differenceadjustment parameter PP₁ to make the intensity of the detected leakedelectromagnetic field small, the leaked electromagnetic field at thepredetermined place can be decreased. Further, by detecting the leakedelectromagnetic field at a plurality of places and adjusting theamplitude adjustment parameters P₁ and P₂ and current phase differenceadjustment parameter PP₁ so as to make the ratio of intensities of thedetected leaked electromagnetic fields a predetermined value, the leakedelectromagnetic field of the desired place can be reduced.

While certain embodiments have been described, these embodiments havebeen presented by way of example only, and are not intended to limit thescope of the inventions. Indeed, the novel embodiments described hereinmay be embodied in a variety of other forms; furthermore, variousomissions, substitutions and changes in the form of the embodimentsdescribed herein may be made without departing from the spirit of theinventions. The accompanying claims and their equivalents are intendedto cover such forms or modifications as would fail within the scope andspirit of the inventions.

1. A drive device driving “N” number (N is an integer of “2” or greater) of inverters corresponding to transmission coil units, the inverters each including a first switching element and a second switching element connected together at respective one ends and a third switching element and a fourth switching element connected together at respective one ends, a connection node of the first and the second switching element being connected to one end of each corresponding transmission coil unit, a connection node of the third and the fourth switching element being connected to another end of each corresponding transmission coil unit, and the inverters each generating AC power by driving the first to fourth switching elements based on a first power-supply voltage supplied to other ends of the first and third switching elements and a second power-supply voltage supplied to other ends of the second and fourth switching elements, and outputting the AC power to each corresponding transmission coil unit, the drive device comprising: a switching signal generation circuit configured to generate switching signals to drive the first to fourth switching elements of each inverter to complementarily drive the first switching element and the second switching element, and complementarily drive the third switching element and the fourth switching element so that a phase difference between an output current of an “M”th (“M” is an integer of 2 or greater and “N” or below) inverter and an output current of an “M−1”th inverter becomes or approach “360×L/N” degrees (“L” is an integer of “1” or greater and less than “N”) and supply the switching signals to the first to fourth switching elements of the inverters.
 2. The drive device according to claim 1, wherein each of the switching signals Is a pulse signal having a same duty ratio and a same frequency.
 3. The drive device according to claim 1, wherein a phase difference between a switching signal of the first switching element in the “M−1”th inverter and a switching signal of the third switching element in the “M−1”th inverter is “180−P_(M−1)” degrees, the phase difference between a switching signal of the first switching element in the “M”th inverter and a switching signal of the third switching element in the “M”th inverter is “180−P_(M)” degrees, and the drive device generates the switching signals so that the phase difference between the switching signal of the first switching element in the “M”th inverter and the switching signal of the first switching element in the “M−1”th inverter becomes or approach “PP_(M−1)−0.5(P_(M−1)−P_(M))” degrees, and the “PP_(M−1)” has a value based on the “360×L/N” degrees.
 4. The drive device according to claim 3, wherein the “PP_(M−1)” has a value obtained by adjusting the “360×L/N” degrees in accordance with a phase lead of an output current relative to an output voltage of the “M”th inverter and a phase lead of an output current relative to an output voltage of a reference inverter selected from the “N” number of inverters.
 5. The drive device according to claim 3, further comprising: a parameter determination circuit configured to acquire information representing a magnitude of an output current or an input current of each inverter from a detection circuit detecting the magnitude of the output current or the input current of each inverter and adjust P₁ to P_(N) so that a difference in magnitude between output currents or input currents of the respective inverters becomes small.
 6. The drive device according to any one of claims 3 to 5, further comprising: a parameter determination circuit configured to obtain information representing a phase of an output current of each inverter from a detection circuit detecting the phase of the output current of each inverter, and to adjust, based on the information obtained from the detection circuit, “PP₁ to PP_(N−1)” so that a phase difference between an output current of the “M”th inverter and an output current of the “M−1”th inverter becomes or approach the “360×L/N” degrees.
 7. The drive device according to claim 3, further comprising: a parameter determination circuit configured to acquire, from a detection circuit detecting an electric field, magnetic Field, or electromagnetic field in a place apart from each transmission coil unit, information representing an intensity of an electric field, magnetic field, or electromagnetic field in the place and adjust at least one of the PP_(M−1), the P_(M−1), or the P_(M) based on the acquired information.
 8. The drive device according to claim 3, further comprising: a parameter determination circuit configured to acquire, from a plurality of detection circuits detecting an electric field, magnetic field, or electromagnetic field in different places apart from each transmission coil unit, information representing an Intensity of an electric field, magnetic field, or electromagnetic field in each of the places and adjust at least one of the PP_(M−1), the P_(M−1), or the P_(M) based on a relation of respective information for the places.
 9. The drive device according to claim 3, further comprising: a parameter determination circuit configured to acquire, from a detection circuit detecting an electric field, magnetic field, or electromagnetic field in a place apart from each transmission coil unit, information representing an intensity of the electric field, magnetic field, or electromagnetic field in the place, and set PP_(M−1) to “0” in a case where a value of the information obtained from the detection circuit is a threshold or less with the “PP_(M−1)” tentatively set to “0”.
 10. The drive device according to claim 1, wherein the first power-supply voltage and the second power-supply voltage are commonly provided to the “N” number of inverters from a same DC power supply.
 11. The drive device according to claim 1, wherein a value of “L” is “1”.
 12. The drive device according to claim 1, wherein a value of “N” is either “2” or “3”.
 13. A wireless power transmission device comprising: a drive device according to claim 1; “N” number of inverters; and “N” number of transmission coil units to receive AC power supply from the “N” number of inverters,
 14. A driving method of driving “N” number (N is an integer of “2” or greater) of inverters corresponding to transmission coil units, the inverters each including a first switching element and a second switching element connected together at respective one ends and a third switching element and a fourth switching element connected together at respective one ends, a connection node of the first and the second switching element being connected to one end of each corresponding transmission coil unit, a connection node of the third and the fourth switching element being connected to another end of each corresponding transmission coil unit, and the inverters each generating AC power by driving the first to fourth switching elements based on a first power-supply voltage supplied to other ends of the first and third switching elements and a second power-supply voltage supplied to other ends of the second and fourth switching elements, and outputting the AC power to each corresponding transmission coil unit, the method comprising: generating switching signals to drive the first to fourth switching elements of each inverter to complementarily drive the first switching element and the second switching element, and complementarily drive the third switching element and the fourth switching element so that a phase difference between an output current of an “M”th (“M” is an integer of 2 or greater and “N” or below) inverter and an output current of an “M−1”th inverter becomes or approach “360×L/N” degrees (“L” is an integer of “1” or greater and less than “N”), each of the switching signals being a pulse signal having a same duty ratio and a same frequency and supplying the switching signals to the first to fourth switching elements of the inverters.
 15. The drive device according to claim 3, wherein the P_(M) is not same for all “M” values. 